Phase difference detection device and method for a position detector

ABSTRACT

Two A.C. output signals amplitude-modulated in accordance with two function values (sine and cosine) differing from each other in correspondence to a position-to-be-detected are received from a position sensor such as a resolver. By performing an addition or subtraction between a signal derived by shifting the electric phase of one of the received A.C. output signals by a predetermined angle, and the other received signal, two electric A.C. signals (sin(ωt±d+θ), sin(ωt±d−θ)) are electrically synthesized which have electric phase angles (θ) corresponding to the position-to-be-detected and are phase-shifted in opposite directions. “±d” here represents phase variation error caused by factors, other than the position-to-be-detected, such as temperature change. In the synthesized two signals, the phase variation errors (±d) appear in the same direction, while the phase differences (θ) corresponding to the position are shifted in opposite, positive and negative, directions. Thus, by measuring the respective phase shift amounts (±d+θ, ±d−θ) and performing appropriate operation, it is allowed to cancel out or extract the error (±d) so that an accurate phase difference (θ) can be detected. Position detection data indicative of the detected phase difference (θ) is converted into a pulse-width-modulated signal and transmitted in the pulse-width-modulated form.

RELATED APPLICATIONS

This application is a continuation-in-part application of ourcorresponding U.S. application Ser. No. 08/818,974 filed Mar. 14, 1997,now U.S. Pat. No. 6,034,624.

BACKGROUND OF THE INVENTION

The present invention relates to a phase difference detection device andmethod for use in position detection and a position detection systemwhich are applicable to detection of both rotational positions andlinear positions, such as a rotational position detector like a resolveror synchro, or a linear position detector based on a similar positiondetecting principle. More particularly, the present invention relates toa technique to detect an absolute position on the basis of an electricphase difference.

Among various induction-type rotational position detectors, one whichproduces two-phase (sine phase and cosine phase) outputs in response toa single-phase exciting input is commonly known as a “resolver”, andothers which produce a three-phase (phases shifted 120° in relation toone another) outputs in response to a single-phase exciting input isknown as a “synchro”. The oldest-fashioned resolvers have double-pole(sine pole and cosine pole) secondary windings provided on the stator insuch a manner to cross each other at a mechanical angle of 90°, with aprimary winding provided on the rotor (the relationship between theprimary and secondary windings may be reversed depending on a desiredapplication). However, the resolvers of this type are disadvantageous inthat they require brushes for electric contact with the primary windingon the rotor. Brushless resolvers eliminating the need for such brushesare also known, where a rotary transformer is provided on the rotor inplace of the brushes.

R/D converters have long been known as a detection system which obtainsposition detection data in digital form by use of a resolver whichproduces a two-phase (sine phase and cosine phase) outputs in responseto a single-phase exciting input.

U.S. Pat. No. 3,648,042 discloses a technique which provides a rotationangle detection signal of a resolver as an analog voltage signal.Further, U.S. Pat. No. 4,011,440 discloses a technique which generates,on the basis of an output signal from a resolver, a cyclic square-wavesignal having a pulse width corresponding to a detected angle andprovides an angular rate on the basis of differences between pulsewidths in individual cycles.

Another detection system is also known, where the resolver excitingmethod is modified to produce a single-phase output in response totwo-phase exciting inputs so that an output signal containing anelectric phase difference angle corresponding to rotational angle θ isobtained to thereby derive digital data indicative of a detected angleθ. Specific examples of the above-mentioned phase difference detectionsystem are disclosed in U.S. Pat. Nos. 4,754,220, 4,297,698, etc.

As known in the art, windings of a sensor such as the resolver tend toundesirably change in impedance under the influence of ambienttemperature change, and thus electric phase of A.C. signals induced in asecondary winding subtly fluctuates in response to the temperaturechange. Additionally, the electric phase of the induced A.C. signalsreceived by a detection circuit varies under the influence of variousfactors other than a position-to-be-detected, such as ununiform wiringlengths between the windings of the sensor and the detection circuit anddelays in various circuit operations. If the phase variation based onthe various factors, other than the position-to-be-detected, such as thetemperature change is expressed by “±d” for convenience of description,in the former-type detection system, i.e., the R/D converter, thevariation amount “±d” is in effect cancelled out and hence has no effectat all on the detecting accuracy. Therefore, it can be seen that thedetection system like the R/D converter is a high-accuracy systeminsusceptible to adverse influence of the ambient temperature change.However, because this detection system is based on a so-called“successive incrementing method” where, as noted earlier, a resettrigger signal is periodically applied to a sequential phase generationcircuit at optional timing to reset a phase angle φ to “0” so as toinitiate incrementing of the angle φ, and the incrementing of the phaseangle φ is stopped upon arrival at “0” of the output of a subtracter tothereby obtain digital data indicative of a detected angle θ, it has towait for a period from the time when the reset trigger is given to thetime when the phase angle φ coincides with the detected angle θ andhence presents poor response characteristics.

On the other hand, in the latter-type detection system, the phasevariation amount “±d” based on the non-positional factors (other thanthe position-to-be-detected) such as temperature change presents a verysignificant problem that the variation “±d” directly appears as adetection error.

The scheme of generating phase detection data in digital representationpermits a high-accuracy detection, but is disadvantageous in that itwould require an increased number of detection-signal transmitting linesif the digital detection data are transmitted directly in a parallelfashion.

SUMMARY OF THE INVENTION

It is therefore an object of the present invention to provide a phasedifference detection device and method for use with a position detectorwhich can perform a high-accuracy position detection without beinginfluenced by unwanted phase variation caused by various factors, otherthan a position-to-be-detected, such as impedance change in a positionsensor due to temperature change, which presents superior high-speedresponse characteristics, or which can significantly simplifydetection-signal transmission lines and also minimize adverse influencesof external disturbances, such as temperature changes, on the detectionsignal on the transmission lines.

In order to accomplish the above-mentioned object, the present inventionprovides a phase difference detection device for a position detector,said position detector being excited by a predetermined reference signalto generate first and second A.C. output signals, said first A C. outputsignal having been amplitude-modulated using, as an amplitudecoefficient, a first function value corresponding to aposition-to-be-detected, and said second A.C. output signal having beenamplitude-modulated using, as an amplitude coefficient, a secondfunction value corresponding to the position-to-be-detected, said phasedifference detection device comprises: a phase shift circuit operativelycoupled to said position detector to shift an electric phase of saidreceived first A.C. output signal by a predetermined angle; a firstcircuit operatively coupled to said phase shift circuit and saidposition detector to perform an operation between an output signal ofsaid phase shift circuit and said second A.C. output signal so as tosynthesize a first data signal having an electric phase angle shifted inone of positive and negative directions in correspondence to theposition-to-be-detected; a second circuit operatively coupled to saidphase shift circuit and said position detector to perform an operationbetween an output signal of said phase shift circuit and said secondA.C. output signal so as to synthesize a second data signal having anelectric phase angle shifted in other of positive and negativedirections in correspondence to the position-to-be-detected; a firstoperation circuit operatively coupled to said first circuit to measurean electric phase difference between said predetermined reference signaland said first data signal to obtain first phase data; a secondoperation circuit operatively coupled to said second circuit to measurean electric phase difference between said predetermined reference signaland said second data signal to obtain second phase data; a thirdoperation circuit operatively coupled to said first and second operationcircuit to calculate position detection data corresponding to theposition-to-be-detected on the basis of said first and second phasedata; and a pulse-width modulation circuit coupled to said thirdoperation circuit to generate a signal pulse-width-modulated inaccordance with the position detection data.

The position detector generates the first and second output signals(e.g., sinθ·sinωt and cosθ·sinωt) amplitude-modulated by two differentfunction values in correspondence to the position-to-be-detected (x) isa known detector or sensor such as a resolver. The present invention ischaracterized in that output signals from such a known position detector(i.e., output signals to which phase-modulation corresponding to theposition-to-be-detected has not been applied) is input to the device fordetection of a phase difference thereof so that an absolute position canbe detected on the basis of the phase difference detection, and thedetected absolute position data is converted to a pulse-width-modulatedsignal for transmission.

Namely, by performing an addition and/or subtraction between the outputsignal (sinθ·cosωt) derived by shifting the electric phase of the firstA.C. output signal received from the position detector by apredetermined angle and the second A.C. output signal (cosθ·sinωt), thefirst and second data signals (e.g., sin(ωt+θ) and sin(ωt−θ)) aresynthesized which have an electric phase angle corresponding to theposition-to-be-detected. In a specific example, the first data signal(sin(ωt+θ)) phase-shifted in a positive direction can be synthesized onthe basis of the addition, e.g., (sinθ·cosωt+cosθ·sinωt), while thesecond data signal (sin(ωt−θ)) phase-shifted in a negative direction canbe synthesized on the basis of the subtraction, e.g.,(−sinθ·cosωt+cosθ·sinωt).

If a fundamental time-varying phase of the obtained A.C. signal isrepresented by “ωt” and phase variation caused by impedance change ofthe detector's wiring due to temperature change and other factors thanthe position-to-be-detected (i.e., non-positional factors) isrepresented by “±d”, then the first data signal may be expressed as“sin(ωt±d+θ)” and the second data signal as “sin(ωt±d−θ)”. That is, theelectrical phase differences (θ), corresponding to theposition-to-be-detected (x), for the first and second data signalsappear as opposite (positive and negative)-direction phase shifts.However, the phase variations “±d” for both of the first and second datasignals have effects in the same positive or negative directiondepending on the current conditions. Thus, by measuring the respectivephase differences “(±d+θ)” and “(±d−θ)” of the first and second datasignals and performing an appropriate operation such as an additionand/or subtraction on the measured differences, it is allowed to cancelout or extract the phase variation “±d” and also detect the phasedifference (θ) free of the phase variation “±d” which accuratelycorresponds to the position-to-be-detected (x).

The detected phase difference (θ), i.e., position detection data, ispulse-with-modulated via a pulse-width modulation circuit, to provide apulse-width-modulated position detection signal having a pulse withcorresponding to the phase difference (θ). The pulse-width-modulatedposition detection signal is then supplied via transmission wirings orlines to another device utilizing the signal (hereinafter called a“utilizing device”). Transmission of the detected position data in theform of the pulse-width-modulated position detection signal isadvantageous in that it can simplify the transmission wirings or linesand prevent voltage level variations of the signal, due to variousadverse influences on the signal passing through the wirings such asimpedance variations by the influence of wiring capacity, noise andtemperature changes, from causing errors to the position detectionsignal, thereby constantly guaranteeing a high detection accuracy.

Consequently, the present invention permits a high-accuracy positiondetection without being influenced by various factors, other than theposition-to-be-detected, such as impedance change of the sensor due totemperature change and ununiform lengths of wiring cables. Further,because the present invention is based on the technique of measuring aphase difference (θ) in A.C. signals, the instant latching method may beemployed rather than the conventional successive incrementing method,and thus the invention can achieve a phase difference detection deviceor method which presents superior high-speed characteristics. Inaddition, by transmitting the detected position data in the form of thepulse-width-modulated position detection signal as mentioned above,there is achieved the superior benefit that the transmission wirings orlines can be significantly simplified and various adverse influences onthe signal passing through the wirings can be reliably avoided.

In one implementation, the third operation circuit may calculate theposition detection data as analog position detection data, and thepulse-width modulation circuit may include an analog circuit forprocessing the analog position detection data.

Alternatively, the third operation circuit may calculate the positiondetection data as digital position detection data, and the pulse-widthmodulation circuit may include a digital circuit for processing thedigital position detection data.

In another alternative, the third operation circuit may calculate theposition detection data as digital position detection data, and thephase difference detection device further may include a converter forconverting the digital position detection data into analog positiondetection data. In this case, the pulse-width modulation circuit mayinclude an analog circuit for processing the analog position detectiondata.

According to another aspect of the present invention, there is provideda phase difference detection device for a position detector, theposition detector being excited by a predetermined reference signal togenerate first and second A.C. output signals, the first A C. outputsignal having been amplitude-modulated using, as an amplitudecoefficient, a first function value corresponding to aposition-to-be-detected, and the second A.C. output signal having beenamplitude-modulated using, as an amplitude coefficient, a secondfunction value corresponding to the position-to-be-detected, which ischaracterized by comprising: a phase shift circuit operatively coupledto the position detector to shift an electric phase of the receivedfirst A.C. output signal by a predetermined angle; a first circuitoperatively coupled to the phase shift circuit and the position detectorto perform an operation between an output signal of the phase shiftcircuit and the second A.C. output signal so as to synthesize a firstdata signal having an electric phase angle shifted in one of positiveand negative directions in correspondence to theposition-to-be-detected; a second circuit operatively coupled to thephase shift circuit and the position detector to perform an operationbetween an output signal of the phase shift circuit and the second A.C.output signal so as to synthesize a second data signal having anelectric phase angle shifted in other of positive and negativedirections in correspondence to the position-to-be-detected; and a thirdcircuit operatively coupled to the first and second circuit to generate,on the basis of a difference between the first data signal and thesecond data signal, a signal pulse-modulated in accordance with positiondata indicative of the position-to-be-detected.

The present invention may be constructed and implemented not only as theabove-mentioned device invention but also as a method invention. Themethod may be arranged and implemented as a program for execution by acomputer, microprocessor or the like, as well as a machine-readablestorage medium storing such a program.

For better understanding of the features of the present invention, thepreferred embodiments of the invention will be described below withreference to the accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

In the accompanying drawings:

FIG. 1 is a perspective view, with parts broken away, of an example of alinear position detector device which is applicable to a phasedifference detection device according to the present invention;

FIG. 2 is a schematic circuit diagram showing a structural example of awinding section of FIG. 1;

FIG. 3 is a block diagram showing an embodiment of the phase differencedetection device according to the present invention;

FIGS. 4A and 4B are diagrams explanatory of the operation of the deviceshown in FIG. 3;

FIG. 5 is a block diagram illustrating a modified embodiment of thephase difference detection device according to the present invention tobe attached to the device shown in FIG. 3;

FIG. 6 is a block diagram illustrating another embodiment of the phasedifference detection device according to the present invention;

FIGS. 7A to 7C are diagrams explanatory of the operation of the deviceshown in FIG. 6;

FIGS. 8A and 8B are block diagrams showing still another embodiment ofthe phase difference detection device according to the presentinvention, where analog position detection data is obtained throughanalog arithmetic operations;

FIG. 9 is a block diagram showing an example of a circuit for measuringand counting pitch-by-pitch displacement of magnetic response members inthe linear position detector device which is applicable to a phasedifference detection device according to the present invention;

FIGS. 10A and 10B are schematic axial and radial sectional views showingan example where positions over a long range beyond one pitch length ofthe magnetic response members are detected in absolute values in thelinear position detector device which is applicable to a phasedifference detection device according to the present invention;

FIG. 11A is a block diagram showing an arrangement where analog positiondetection data output from an analog phase detection circuit, similar tothe one shown in FIG. 8A or 8B, is transmitted after being subjected topulse width modulation;

FIG. 11B is a waveform diagram showing how the pulse@width modulation isperformed in the arrangement of FIG. 11A;

FIG. 11C is a timing chart explanatory of exemplary procedures by whicha utilizing device of FIG. 11 reproduces the position detection data;

FIG. 12A is a block diagram showing an arrangement where digitalposition detection data output from a digital phase detection circuit,similar to the one shown in FIG. 3, 5 or 6, is subjected to pulse widthmodulation;

FIG. 12B is a block diagram showing another arrangement where digitalposition detection data output from a digital phase detection circuit,similar to the one shown in FIG. 3, 5 or 6, is subjected to pulse widthmodulation; and

FIG. 13 is a block diagram showing another arrangement where a signalformed by pulse-width modulating the position detection data isgenerated on the basis of two zero-cross detection pulses output fromthe zero-cross detection circuit of FIG. 3 using the same phasedifference detection circuit as shown in FIG. 8B.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

FIG. 1 is a perspective view of an example of a linear position detectordevice which is applicable to a phase difference detection deviceaccording to the present invention. The linear position detector devicegenerally comprises a winding section 10 and a variable magneticcoupling section 20. The variable magnetic coupling section 20, which iscoupled to a predetermined mechanical system (not shown) that is anobject of detection by the detector device, is capable of linearlyreciprocating in response to a varying linear position of the mechanicalsystem. On the other hand, the winding section 10 is positionally fixedin a suitable manner. Thus, the variable magnetic coupling section 20linearly moves relative to the winding section 10, in response to avarying linear position of the mechanical system to be detected (objectof detection). Conversely, the winding section 10 may be constructed tomove in response to a varying linear position of the mechanical systemto be detected, with the variable magnetic coupling section 20 fixed inposition. In short, this detector device is constructed to detect alinear position of the variable magnetic coupling section 20 relative tothe winding section 10. The direction of such a relative lineardisplacement is denoted in FIG. 1 by a double-head arrow X.

The winding section 10 includes primary windings PW1 to PW5 which areexcited by a common single-phase A.C. signal, and secondary windings SW1to SW4 provided at different locations with respect to the lineardisplacement direction X. The winding section 10 is shown in partialcross section in FIG. 1 to clearly illustrate the structuralrelationships between the first and second windings; actually, thewinding coils of the winding section 10 are disposed on the rod-shapedvariable magnetic coupling section 20 with an appropriate gap lefttherebetween as additionally denoted by dotted line. Because the primarywindings PW1 to PW5 are excited by the common single-phase A.C. signalin the instant embodiment, either an integrally formed single winding ora predetermined plurality of discrete windings may be arranged in anysuitable manner. However, it is preferable that the predeterminedprimary windings PW1 to PW5 be arranged in such a manner that each ofthe secondary windings SW1 to SW4 is interposed between adjacent primarywindings PW1 to PW5, because magnetic fields generated by the primarywindings can effectively operate on or influence the individualsecondary windings SW1 to SW4 and later-described magnetic responsemembers 22 of the variable magnetic coupling section 20 can effectivelyinfluence the magnetic fields.

The linear or rod-shaped variable magnetic coupling section 20 includesa base rod section 21, on which a plurality of the magnetic responsemembers 22 having a predetermined magnetic response characteristic areprovided at a predetermined pitch p along the linear displacementdirection X. As already known in the art, the magnetic response members22 may be made of any suitable material such as a magnetic material likeiron or nickel, or non-magnetic, electrically conductive material likecopper or aluminum, in such a manner that they assume a predeterminedmagnetic response characteristic such as in magnetic permeability,reluctance or eddy-current loss. The base rod section 21 may also bemade of any suitable material such as a magnetic material, non-magneticmaterial or electrically conductive material, depending on a particularmaterial and/or shape of the magnetic response members 22. In otherwords, it is only sufficient that magnetic response characteristicsinfluencing the winding section 10 differ between the place where themagnetic response member 22 is present and the place where the magneticresponse member 22 is not present. The formation, of the magneticresponse members 22, on the rod section 21 may be done by any suitableknown method, such as pasting, adhesive bonding, caulking, cutting,plating, vacuum evaporation and baking. The rod section 21 may be madeof a flexible material such as flexible wire, rather than a rigidmaterial.

As the magnetic response members 22 of the variable magnetic couplingsection 20 change their positions relative to the winding section 10 inresponse to a varying linear position of the object of detection,magnetic coupling between the primary windings PW1 to PW5 and thesecondary windings SW1 to SW4 are also changed in response to thevarying linear position of the object of detection. Consequently,inductive A.C. output signals amplitude-modulated in accordance with thelinear position of the object of detection occur are produced in thesecondary windings SW1 to SW4, with amplitude function characteristicsdiffering depending on the respective locations of the secondarywindings SW1 to SW4. Because the primary windings PW1 to PW5 are excitedby a single-phase A.C. signal, the inductive A.C. output signalsoccurring in the secondary windings SW1 to SW4 are identical inelectrical phase and each of their amplitude functions periodicallychanges in such a manner that a displacement amount corresponding to onepitch length p between the magnetic response members 22 represents onecycle of the periodical change.

The four secondary windings SW1 to SW4 are disposed at predeterminedintervals within a range of one pitch length p of the magnetic responsemembers 22, and set in such a manner that the inductive A.C. outputsignals produced in the individual secondary windings SW1 to SW4 presentdesired amplitude function characteristics. For example, if the detectordevice is constructed as a resolver-type position detector, theamplitude function characteristics of the inductive A.C. output signalsproduced in the individual secondary windings SW1 to SW4 are set torepresent a sine function, cosine function, minus sine function andminus cosine function, respectively. For example, as shown in FIG. 1,the range of one pitch length p is divided into four segments, and thesecondary windings SW1 to SW4 are positioned in the four segmentsdisplaced from each other by an amount “p/4”. By so doing, the amplitudefunction characteristics of the inductive A.C. output signals producedin the individual secondary windings SW1 to SW4 are set to represent asine function, cosine function, minus sine function and minus cosinefunction, respectively. Of course, the respective locations of theindividual windings can be varied subtly depending on variousconditions; thus, the embodiment is designed so as to ultimately obtaindesired amplitude function characteristics by adjusting the locations ofthe individual windings or by adjusting secondary output levels throughelectrical amplification.

In the situation where the output from the secondary winding SW1represents a sine function (denoted by “s” in the figure), the secondarywinding SW3 displaced from the winding SW1 by an amount of “p/2”provides an output representing a minus sine function /s (the mark “/”in the text corresponds to the upper short bar in the figure); in thiscase, a first A.C. output signal having a sine amplitude function isprovided by differentially synthesizing the two outputs. Similarly, thesecondary winding SW2 displaced from the winding SW1, representing thesine function output, by an amount of “p/4” provides an outputrepresenting a cosine function (denoted by “c” in the figure), and thesecondary winding SW4 displaced from the winding SW1 by an amount of“p/2” provides an output representing a minus cosine function /c (themark “/” in the text corresponds to the upper short bar in the figure);in this case, a second A.C. output signal having a cosine amplitudefunction is provided by differentially synthesizing the two outputs.

FIG. 2 is a schematic circuit diagram of the winding section 10, inwhich a common exciting A.C. signal (denoted by “sinωt”, for convenienceof illustration) is applied to the primary windings PW1 to PW5. Inresponse to excitation of the primary windings PW1 to PW5, A.C. signalshaving amplitude values corresponding to locations of the magneticresponse members 22 relative to the winding section 10 are induced inthe individual secondary windings SW1 to SW4. The induced voltage levelsrepresent two-phase function characteristics of sinθ and cosθ and twoopposite-phase function characteristics of −sinθ and −cosθ, incorrespondence with a current linear position of the object of detectionx. That is, the inductive output signals of the individual secondarywindings SW1 to SW4 are amplitude-modulated by the two-phase functioncharacteristics of sinθ and cosθ and two opposite-phase functioncharacteristics of −sinθ and −cosθ in correspondence with a currentlinear position of the object of detection. Note that “θ” isproportional to “x”, and, for example, θ=2π(x/p). For convenience ofexplanation, coefficients, such as the respective numbers of turns ofthe windings, depending on other conditions are not considered here.Also, the secondary winding SW1 is shown and described as a sine phasewith its output signal represented as “sinθ·sinωt”; the secondarywinding SW2 is shown and described as a cosine phase with its outputsignal represented as “cosθ·sinωt”; the secondary winding SW3 is shownand described as a minus sine phase with its output signal representedas “−sinθ·sinωt”; and the secondary winding SW4 is shown and describedas a minus cosine phase with its output signal represented as“−cosθ·sinωt”. By differentially synthesizing the inductive outputs ofthe sine and minus sine phases, there will be obtained the first A.C.output signal (2sinθ·sinθt) having a sine amplitude function. Similarly,by differentially synthesizing the inductive outputs of the cosine andminus cosine phases, there will be obtained the second A.C. outputsignal (2cosθ·sinωt) having a cosine amplitude function. Hereinafter,the coefficient “2” will be omitted for simplicity of illustration, sothat the first A.C. output signal will be indicated as “sinθ·sinωt” andthe second A.C. output signal will be indicated as “cosθ·sinωt”.

In the above-mentioned manner, there are provided the first A.C. outputsignal A(=sinθ·sinωt) having, as its amplitude value, a first functionvalue sinθ corresponding to the linear position of the object ofdetection x and the second A.C. output signa B(=cosθ·sinωt) having, asits amplitude value, a second function value cosθ corresponding to thesame linear position of the object of detection x. It will be seen thatwith such winding arrangements, the linear position detector is capableof providing two A.C. output signals having two-phase amplitudefunctions (sine and cosine outputs) just like those provided by theconventional rotary-type position detector devices commonly known asresolvers. As a result, the two-phase A.C. output signals (A=sinθ·sinωtand B=cosθsinωt) can be utilized in a similar manner to the outputs fromthe conventionally known resolvers.

A position detector applicable to a phase difference detection deviceaccording to the present invention is not limited to the positiondetector device as shown in FIG. 1. The position detector applicable toa phase difference detection device according to the present inventionmay be a position sensor of any desired type having a single-phaseexciting input and two-phase outputs. For example, the position detectormay be a conventionally known resolver of the brushless orbrush-equipped type. Alternatively, the position detector may be avariable-reluctance type position sensor such as a “Microsyn” (tradename) where primary and secondary windings are provided on the statorwith no winding on the rotor or movable member, or it may be either arotational position sensor or linear position detecting sensor. Theposition detector applicable to the phase diference detection deviceaccording to the present invention may be constructed in such a mannerthat the winding section has only a primary wninding or windings and anoutput signal or signals is generated from the primary winding orwindings in response to an impedance or inductance variation of theprimary winding or windings.

FIG. 3 shows an embodiment of the phase difference detection deviceaccording to the present invention. A single-phase exciting A.C. signal(denoted by “sinωt” for convenience of description) generated in adetection circuit section 11 is applied to the winding section 10 so asto excite the primary winding. In the winding section 10, A.C. outputsignals are induced in the two-phase secondary windings in response toexcitation of the primary winding W1, and the respective induced voltagelevels of the signals present two-phase functional characteristics, sinθand cosθ, corresponding to a particular position to be detected(position-to-be-detected) x. That is, the induced output signals of thesecondary windings are output with their amplitudes modulated by thetwo-phase functional characteristics sinθ and cosθ corresponding to theposition-to-be-detected x. For convenience of description, coefficientsrelating to other conditions such as the respective turns of thewindings will not be, taken into account, and the secondary winding isassumed to be of sine phase with its output signal represented by“sinθ·sinωt” whereas the other secondary winding is assumed to be ofcosine phase with its output signal represented by “cosθ·sinωt”. Thatis, the secondary winding outputs a first A.C. output signalA(=sinθ·sinωt) having as its amplitude value a first function value sinθcorresponding to the position-to-be-detected x, and the secondarywinding outputs a second A.C. output signal B(=cosθ·sinωt) having as itsamplitude value a second function value cosθ corresponding to theposition-to-be-detected x.

In the detection circuit section 41 shown in FIG. 3, counter 42 countspredetermined high-speed clock pulses CK, exciting signal generationcircuit 43 generates an exciting A.C. signal (e.g., sinωt) on the basisof a counted value of the counter 42, and the generated A.C. signal issupplied to the primary winding W1 of the winding section 10. Themodulus of the counter 42 corresponds to one cycle of the exciting A.C.signal, and it is assumed herein, for convenience of description, thatits counted value “0” corresponds to the zero phase of reference sinesignal sinωt. During one complete cycle of the reference sine signalsinωt from the zero to maximum phases is generated during one cycle ofcounting of the counter 42 from zero to the maximum value, one completecycle of the exciting A.C. signal sinωt is generated, by the excitingsignal generation circuit 43.

The first and second A.C. output signals A and B of the winding section10 are supplied to the detection circuit section 41. In the detectioncircuit section 41, the first A.C. output signal A(=sinθ·sinωt) is inputto a phase shift circuit 44 so that it is shifted in electric phase by apredetermined amount (e.g., 90°) so as to provide a phase-shifted A.C.signal A′(=sinθ·cosωt). The detection circuit section 41 also includesadder and subtracter circuits 45 and 46. In the adder circuit 45, thephase-shifted A.C. signal A′(=sinθ·cosωt) from the phase shift circuit44 and the above-mentioned second A.C. output signal B(=cosθ·sinωt) areadded together so as to obtain, as an added output signal, a firstelectric A.C. signal Y1 that may be expressed by a brief formula ofB+A′=cosθ·sinθt+sinθ·cosωt=sin(ωt+θ). On the other hand, in thesubtracter circuit 46, a subtraction between the phase-shifted A.C.signal A′(=sinθ·cosωt) from the phase shift circuit 44 and theabove-mentioned second A.C. output signal B(=cosθ·sinωt) is performed soas to obtain, as a subtracted output signal, a second electric A.C.signal Y2 that may be expressed by a brief formula ofB−A′=cosθ·sinωt−sinθ·cosωt=sin(ωt−θ). In this way, there can beobtained, through electric processing, the first electric A.C. signalY1(=sin(ωt+θ)) having an electric phase (+θ) shifted in the positivedirection in correspondence to the position-to-be-detected x, and thesecond electric A.C. signal Y2(=sin(ωt−θ)) having an electric phase (−θ)shifted in the negative direction in correspondence to theposition-to-be-detected x.

The above-mentioned output signals Y1 and Y2 of the adder and subtractercircuits 45 and 46 are given to zero-cross detection circuits 47 and 48for detection of the respective zero-cross points of the signals Y1 andY2. The zero-cross detection is done by, for example, identifying apoint where the signal Y1 or Y2 changes from a negative value to apositive value, i.e., a zero phase point. Zero-cross detection pulsesgenerated by the circuits 47 and 48 upon detection of the respectivezero-cross points are applied as latch pulses LP1 and LP2 tocorresponding latch circuits 49 and 50. Each of the latch circuits 49and 50 latches a counted value of the counter 42 at the timing of thecorresponding latch pulse LP1 or LP2. Since, as noted earlier, themodulus of the counter 42 corresponds to one cycle of the exciting A.C.signal and its counted value “0” corresponds to a zero phase of thereference sine signal sinωt, data D1 and D2 thus latched in the latchcircuits 49 and 50 correspond to phase differences of the output signalsY1 and Y2 with respect to the reference sine signal sinωt. Output datafrom the latch circuits 49 and 50 are supplied to an error calculationcircuit 51, which in turn conducts a computation of “(D1+D2)/2”. Thiscomputation may in practice be conducted by right (downward)-shifting byone bit the sum of the binary data “D1+D2”.

If the phase variation error is represented by “±d” considering possibleinfluence of ununiform lengths of wiring cables between the windingsection 10 and detection circuit section 41 and impedance change causedby the temperature change in the windings of the winding section 10, theabove-mentioned signals handled in the detection circuit section 41 maybe expressed as follows:

A=sin θ·sin(ωt±d);

A′=sin θ·cos(ωt±d);

B=cos θ·sin(ωt±d);

Y 1=sin(ωt±d+θ);

Y 2=sin(ωt±d−θ);

D 1=±d+θ; and

D 2=±d−θ

Namely, since the phase difference counting is performed using thereference sine signal sinθt as a reference phase, the phase differencemeasurement data D1 and D2 will contain the phase variation error “±d”as previously mentioned. The phase variation error “±d” can becalculated by the error calculation circuit 51 using the followingexpression: $\begin{matrix}{{\left( {{D1} + {D2}} \right)/2} = \quad {\left\{ {\left( {{\pm d} + \theta} \right) + \left( {{\pm d} - \theta} \right)} \right\}/2}} \\{= \quad {{{\pm 2}{d/2}} = {\pm d}}}\end{matrix}$

Data indicative of the phase variation error “±d” calculated by theerror calculation circuit 51 is delivered to a subtracter circuit 52,where the data “±d” is subtracted from one (D1) of the phase differencemeasurement data D1 and D2. That is, because the subtracter circuit 52carries out a subtraction of “D1−(±d)”,

D 1−(±d)=±d+θ−(±d)=θ,

and thus there can be obtained digital data indicative of an accuratephase difference θ from which the phase variation error “±d” has beenremoved. From the foregoing, it will be readily understood that thepresent invention allows only the accurate phase difference θcorresponding to the position-to-be-detected x to be extracted bycancelling out the phase fluctuation error “±d”.

This feature will be described in greater detail with reference to FIGS.4A and 4B, which show waveforms, at and around a zero phase point, ofthe sine signal sinωt used as the phase measuring reference and thefirst and second A.C. signals Y1 and Y2; FIG. 4A shows such waveforms inthe case where the phase variation error is positive, whereas FIG. 4Bshows such waveforms in the case where the phase variation error isnegative. In the case shown in FIG. 4A, the zero phase of the firstsignal Y1 is displaced or shifted, by “θ+d”, ahead of that of thereference sine signal sinωt, and phase difference detection data D1corresponding thereto represents a phase difference equivalent to “θ+d”.Further, the zero phase of the second signal Y2 is displaced or shifted,by “−θ+d”, behind that of the reference sine signal sinωt, and phasedifference detection data D2 corresponding thereto represents a phasedifference equivalent to “−θ+d”. In this case, the error calculationcircuit 51 calculates a phase variation error “+d” on the basis of$\begin{matrix}{{\left( {{D1} + {D2}} \right)/2} = \quad {\left\{ {\left( {{+ d} + \theta} \right) + \left( {{+ d} - \theta} \right)} \right\}/2}} \\{= \quad {{{+ 2}{d/2}} = {+ d}}}\end{matrix}$

Then, the subtracter circuit 52 carries out a calculation of

D 1−(+d)=+d+θ−(+d)=θ,

to thereby extract an accurate phase difference θ.

On the other hand, in the case shown in FIG. 4B, the zero phase of thefirst signal Y1 leads, by “θ−d”, that of the reference sine signalsinωt, and phase difference detection data D1 corresponding theretorepresents a phase difference equivalent to “θ−d”. Further, the zerophase of the second signal Y2 lags, by “−θ−d”, that of the referencesine signal sinωt, and phase difference detection data D2 correspondingthereto represents a phase difference equivalent to “−θ−d”. In thiscase, the error calculation circuit 51 calculates a phase fluctuationerror “+d” on the basis of $\begin{matrix}{{\left( {{D1} + {D2}} \right)/2} = \quad {\left\{ {\left( {{- d} + \theta} \right) + \left( {{- d} - \theta} \right)} \right\}/2}} \\{= \quad {{{- 2}{d/2}} = {- d}}}\end{matrix}$

Then, the subtracter circuit 52 carries out a calculation of

D 1−(−d)=−d+θ−(−d)=θ,

to thereby extract an accurate phase difference θ.

Alternatively, the subtracter circuit 52 may carry out a subtraction of“D2−(±d)”, and by so doing, there can be obtained data (−θ) which inprinciple reflects an accurate phase difference θ in a similar manner tothe above-mentioned.

As seen from FIGS. 4A and 4B as well, the electric phase differencebetween the first and second signals Y1 and Y2 is 2θ, which alwaysrepresents the double of the accurate phase difference θ where the phasevariation errors “±d” in the two signals Y1 and Y2 have been cancelledout. Therefore, the structure of the circuitry including the latchcircuits 49 and 50, error calculation circuit 51, subtracter circuit 52etc. may be modified, if necessary, in such a manner to directly obtainthe electric phase difference 2θ between the first and second signals Y1and Y2. For example, digital data corresponding to the electric phasedifference 2θ where the phase variation errors “±d” in the two signalsY1 and Y2 have been cancelled out may be obtained by using a suitablemeans to gate a period between generation of the pulse LP1 correspondingto a zero phase of the first signal Y1 output from the zero-crossdetection circuit 47 and generation of the pulse LP2 corresponding to azero phase of the second signal Y2 output from the zero-cross detectioncircuit 48, and counting the gated period. Then, data corresponding to θcan be obtained by downward-shifting the digital data by one bit.

The latch circuit 49 for latching “+θ” and latch circuit 50 for latching“−θ” in the above-mentioned embodiment have just been described aslatching a count output of the same counter 42, and no specificreference has been made to the sign (positive or negative sign) of thelatched data. However, the sign of the data may be selected as desiredby applying an appropriate design choice along the spirit of the presentinvention. If, for example, the modulus of the counter 42 is 4,096 (indecimal notation), it will suffice to perform necessary arithmetic byrelating its possible digital counts 0 to 4,095 to phase angles 0 to360°. In the simplest example, the necessary arithmetic may be performedby using the uppermost bit of a counted output of the counter 42 as asign bit and relating digital counts 0 to 2,047 to +0 to +180° anddigital counts 2,048 to 4,095 to −180 to −0°. In another example,digital counts 4,095 to 0 may be related to negative angle data −360 to−0° by the input or output data of the latch circuit 50 into 2'scomplements.

Incidentally, no particular problem arises when theposition-to-be-detected x is in a stationary state; however, as theposition x varies timewise, the corresponding phase angle θ alsotime-varies. In such a case, the phase difference value θ between therespective output signals Y1 and Y2 of the adder and subtracter circuits45 and 46, rather than assuming a fixed value, presents dynamiccharacteristics time-varying in correspondence with the moving speed. Ifthis is represented by θ(t), then the respective output signals Y1 andY2 may be expressed by

Y 1=sin{ωt±d+θ(t)}

Y 2=sin{ωt±d−θ(t)}

Namely, the phase-leading output signal Y1 shifts in frequency, withrespect to the frequency of the reference signal sinωt, in a directionwhere the frequency increases in accordance with the “+θ(t)”, whereasthe phase-lagging output signal Y2 shifts in frequency, with respect tothe frequency of the reference signal sinωt, in a direction where thefrequency decreases in accordance with the “−θ(t)”. Because, under suchdynamic characteristics, the respective periods of the signals Y1 and Y2successively shift in the opposite directions for each cycle of thereference signal sinωt, the measured time references of the latched dataD1 and D2 in the latch circuits 49 and 50 will differ from each other,so that the accurate phase variation errors “±d” can not be obtained bymere operations of the circuits 51 and 52.

A simplest possible way to avoid such a problem is to limit the functionof the device of FIG. 3 in such a manner that the device ignores outputsobtained when the position-to-be-detected x is moving timewise andinstead measures the position x in a stationary state by use of onlyoutputs obtained in the stationary state. Thus, the present inventionmay be embodied for such a limited purpose.

But, it will be desirable to be able to accurately detect every phasedifference θ corresponding to a varying position-to-be-detected x evenduring the time-variation of the object. Therefore, a description willbe made below, with reference to FIG. 5, about an improvement of thepresent invention which, in order to address the above-mentionedproblem, is capable of detecting every phase difference θ correspondingto a varying position x even during the time-variation of the positionx.

FIG. 5 extractively shows a modification of the error calculation andsubtracter circuits 51 and 52 in the detection circuit section 41 ofFIG. 3, and the other components not shown in the figure may be the sameas in FIG. 3. If phase difference θ corresponding to the time-varyingposition-to-be-detected x is represented by +θ(t) and −θ(t), the outputsignals Y1 and Y2 can be expressed as the above-mentioned. Then, thephase difference measurement data D1 and D2 obtained by the latchcircuits 49 and 50 are

D 1=±d+θ(t)

D 2=±d−θ(t)

In this case, “±d+θ(t)” will repeatedly time-vary in the positivedirection over a range from 0 to 360° in response to the time-variationof the phase difference θ, whereas “±d−θ(t)” will repeatedly time-varyin the negative direction over a range from 360 to 0° in response to thetime-variation of the phase difference θ. Thus, although ±d+θ(t)≠±d−θ(t)results sometimes, the variations of the two data intersect each othersome other time, and thereby ±d+θ(t)=±d−θ(t) is established. When±d+θ(t)=±d−θ(t), the output signals Y1 and Y2 are in phase and the latchpulses LP1 and LP2 corresponding to the respective zero-cross detectiontiming of the signals Y1 and Y2 are generated at the same timing.

In FIG. 5, a coincidence detection circuit 53 detects a coincidence inthe generation timing of the latch pulses LP1 and LP2 corresponding tothe respective zero-cross detection timing of the output signals Y1 andY2, and generates a coincidence detection pulse EQP upon detection ofsuch a coincidence. A time-variation determination circuit 54determines, via an optional means (e.g., means for detecting presence orabsence of time-variation in the value of one of the phase differencemeasurement data D1), that the position-to-be-detected x is in thetime-varying mode, and it outputs a time-varying mode signal TM uponsuch a detection.

Selector 55 is provided between the error calculation and subtractercircuits 51 and 52 so that when no time-varying mode signal TM isgenerated (TM=“0”), i.e., when the position-to-be-detected x is nottime-varying, the output signal applied from the error calculationcircuit 51 to selector input B is selected to be fed to the subtractercircuit 52. When the input B of the selector 55 is selected, thecircuitry of FIG. 5 operates in a manner equivalent to the circuitry ofFIG. 15; that is, when the position-to-be-detected x is at rest, i.e,not moving, the output data of the calculation circuit 51 is feddirectly to the subtracter circuit 52 via the input B so that thecircuitry operates as in FIG. 3.

In contrast, when the time-varying mode signal TM is generated (TM=“1”),i.e., when the position-to-be-detected x is time-varying, the outputsignal applied from the latch circuit 56 to selector input A is selectedto be fed to the subtracter circuit 52. Then, once the coincidencedetection pulse EQP is generated while the mode signal is “1”, an ANDcondition is satisfied in AND gate 57, which thus outputs a pulseresponsive to the coincidence detection pulse EQP. The output pulse ofthe AND gate 57 is given as a latch command to the latch circuit 56,which latches output count data of the counter 42 in response to thelatch command. Because, when the coincidence detection pulse EQP isgenerated, the output of the counter 42 will be latched concurrently inboth the latch circuits 49 and 50, D1=D2 is met, and hence the datalatched in the latch data 56 is equivalent to D1 or D2 (provided thatD1=D2).

Further, because the coincidence detection pulse EQP is generated oncethe respective zero-cross detection timing of the output signals Y1 andY2 coincides, i.e., once “±d+θ(t)=±d−θ(t)” is met, the data latched inthe latch data 56 in response to the pulse EQP is equivalent to D1 or D2(provided that D1=D2) and therefore equivalent to

(D1+D2)/2

This means $\begin{matrix}{{\left( {{D1} + {D2}} \right)/2} = \quad \left\lbrack {\left\{ {{\pm d} + {\theta (t)}} \right\} + {\left\{ \left( {{\pm d} - {\theta (t)}} \right\} \right\rbrack/2}} \right.} \\{= \quad {{2{\left( {\pm d} \right)/2}} = {\pm d}}}\end{matrix}$

and hence further means that the data latched in the latch data 56 is anaccurate indication of the phase variation error “±d”.

Thus, when the position-to-be-detected x is time-varying, dataaccurately indicating the phase variation error “±d” is latched in thelatch circuit 56 in response to the coincidence detection pulse EQP, andthe output data of this latch circuit 56 is sent via the input A to thesubtracter circuit 52. Accordingly, the subtracter circuit 52 can obtainonly the data θ (θ(t) in the case where the position x is time-varying)which accurately corresponds only to the position x and from which thephase variation error “±d” has been eliminated.

In the modified example of FIG. 5, the AND gate 57 may be omitted sothat the coincidence detection pulse EQP is applied directly to thelatch control input of the latch circuit 56.

Further, as denoted by a broken-line arrow, the latch circuit 56 maylatch the output data “±d” of the error calculation circuit 51 ratherthan the output count data of the counter 42. In such a case, the outputtiming from the calculation circuit 51 of the output data is slightlydelayed behind the generation timing of the coincidence detection pulseEQP due to operational delays of the latch circuits 49 and 50 andcalculation circuit 51, so it is preferable that the output of thecalculation circuit 51 be latched into the latch circuit 56 after havingundergone appropriate adjustment against the time delay.

It should also be readily understood that in the case where thedetection circuit section 41 is constructed taking only dynamiccharacteristics into account, it is possible to omit the circuit 51 andselector 55 of FIG. 5 and one of the latch circuits 49 or 50 of FIG. 3.

FIG. 6 shows another embodiment of the phase difference detectingoperation directed to cancelling phase variation error “±d”.

First and second A.C. output signals A and B which are outputted fromthe secondary winding SW1-SW4 of the winding section 10 are introducedinto a detection circuit section 60. In a same manner as shown in FIG.3, the first A.C. output signal A(=sinθ·sinωt) is input to a phase shiftcircuit 44 of the section 60, where its electric phase is shifted by apredetermined amount to provide a phase-shifted A.C. signalA′(=sinθ·cosωt). In a subtracter circuit 46, a subtraction between thephase-shifted A.C. signal A′(=sinθ·cosωt) and the second A.C. outputsignal B(=cosθ·sinωt) is performed to provide an A.C. signal Y2 that maybe expressed by a brief formula of B−A′=cosθ·sinωt−sinθ·cosωt=sin(ωt−θ).The output signal Y2 of the subtracter circuit 46 is fed to a zero-crossdetection circuit 48 so that a latch pulse LP2 is output upon detectionof a zero-cross point and supplied to a latch circuit 50.

The embodiment of FIG. 6 is different from that of FIG. 3 in terms of areference phase that is used to measure a phase difference amount θ froman A.C. signal Y2(=sin(ωt−θ)) containing the phase differencecorresponding to the position-to-be-detected x. More specifically, inthe embodiment of FIG. 3, the reference phase used to measure the phasedifference amount θ is the zero phase of the reference sine signal sinωtwhich is not input to the winding section 10 and hence does not containphase variation error “±d” caused by various factors such as variationof wiring impedance due to temperature change etc. Because of this, theembodiment of FIG. 3 forms two A.C. signals, Y1(=sin(ωt+θ)) andY2(=sin(ωt−θ)) and cancels out the phase variation error “±d” bycalculating a phase difference between the two signals. In contrast, theembodiment of FIG. 6 is designed to eliminate the phase variation error“±d” by, on the basis of the first and second output signals A and Boutput from the winding section 10, forming the reference phase to beused for measuring the phase difference amount θ in such a manner thatthe reference phase itself contains the error “±d”.

More specifically, in the detection circuit section 60 of FIG. 6, thefirst and second output signals A and B output from the winding section10 are input to zero-cross detection circuits 61 and 62, respectively,each of which detects a zero-cross of the corresponding input signal. Itis assumed herein that each of the detection circuits 61 and 62 outputsa zero-cross detection pulse in response to both a positive-goingzero-cross point where the amplitude of the corresponding input signal Aor B changes from a negative value to a positive value (so to speak, 0°phase) and a negative-going zero-cross point where the amplitude of thecorresponding input signal A or B changes from a positive value to anegative value (so to speak, 180° phase). The reason is that, becausesinθ and cosθ determining the positive or negative polarity of theamplitude of each signal A and B become positive or negative in responseto the value of θ, it is at least necessary to detect a zero-cross forevery 180° in order to detect zero-cross points for every 360° on thebasis of combination of the two signals. The zero-cross detection pulsesoutput from the two zero-cross detection circuits 61 and 62 are ORed byan OR circuit 63, and the resultant output of the OR circuit 63 is fedto a suitable ½ frequency divider/pulse circuit 64 (which may includefor example a ½ frequency divider circuit such as a T flip-flop and apulse outputting AND gate) in such a manner that every other zero-crossdetection pulse is taken out, so that the zero-cross for every 360°,i.e, zero-cross detection pulse corresponding only to the zero phase isoutput as a reference phase signal pulse RP. This pulse RP is applied tothe reset input of a counter 65 which continually counts predeterminedclock pulses CK. The counter 65 is reset to “0” whenever the referencephase signal pulse RP is applied thereto. The counted value of thecounter 65 is fed to the latch circuit 50, where it is latched at thegeneration timing of the latch pulse LP2. Then, the data D thus latchedin the latch circuit 50 is output as measurement data of the phasedifference θ corresponding to the position-to-be-detected x.

The first and second A.C. output signals A and B from the windingsection 10 are expressed by A=sinθ·sinωt and B=cosθ·sinωt, respectively,and are in phase with each other. Respective zero-cross points shouldtherefore be detected at the same timing; actually, however, theamplitude level of either of the signals may become “0” or close to “0”since the amplitude coefficients vary in sinθ and cosθ, in which case itis practically impossible to detect any zero-cross point of one of thesignals. Thus, this embodiment is characterized in that zero-crossdetection processing is performed on each of the two A.C. output signalsA(=sinθ·sinωt) and B(=cosθ·sinωt), and the zero-cross detection outputsof the two signals are ORed so that even when no zero-cross of either ofthe signal can be detected because of a small amplitude level, it ispossible to utilize the zero-cross detection output signal of the othersignal having a relatively great amplitude level.

In the FIG. 6 embodiment, if the phase variation caused by variation inwiring impedance of the winding section 10 etc. is for example “−d”, theA.C. signal Y2 output from the subtracter circuit 46 will be sin(ωt−d−θ)as shown in FIG. 7A. In this case, the output signals A and B of thewinding section 10 assume respective amplitude values sinθ and cosθcorresponding to the angle θ and contain respective phase variationerrors as represented by A=sinθ·sin(ωt−d) and B=cosθ·sin(ωt−d), as shownin FIG. 7B. Consequently, the reference phase signal RP obtained at suchtiming as shown in FIG. 7C on the basis of the zero-cross detection isdisplaced, by the variation error “−d”, from a zero phase of the normalreference reference sine signal sinωt. Thus, an accurate angle value θfree of the variation error “−d” will be obtained by measuring a phasedifference amount in the output A.C. signal Y2(=sin(ωt−d−θ)) of thesubtracter circuit 46.

Note that after various conditions, such as the wiring length of thewinding section 10, have been set, the impedance variation dependsprimarily on the temperature. Then, the above-mentioned phase variationerror ±d corresponds to data indicative of a temperature in anenvironment where the linear position detector device is installed.Thus, the device including the circuit 51 for calculating a phasevariation error ±d as in the embodiment of FIG. 3 can provide thecalculated phase variation error ±d as temperature detection data ifnecessary. As a result, the arrangements of the present inventionaffords the superior benefit that it can not only detect a currentposition of the object of detection but also provide data indicative ofan environmental temperature, using only one position detector, thusachieving a multi-purpose sensor that has not existed so far. Of course,the present inventive arrangements permit a high-precision positiondetection accurately responding to the object of detection, withoutbeing significantly influenced by the sensor impedance variations due totemperature changes and the lengths of wiring cables. Further, becausethe examples of FIGS. 3 and 6 are based on measurement of a phasedifference in A.C. signals, they can provide a detection with higherresponsiveness than that provided by the technique of the prior art.

Whereas the phase data D1 and D2 of the output signals Y1 and Y2, in theforegoing example, have been described as being subjected to digitaloperations so that the position detection data θ is output in digitalrepresentation, the position detection data θ may alternatively beoutput in analog representation. To this end, it is only necessary thatthe calculated position detection data θ undergo D/A (digital-to-analog)conversion. As another example, analog operations may be carried out, bycircuitry as illustrated in FIG. 8A, to directly obtain the positiondetection data θ in analog representation. In the circuitry of FIG. 8A,a zero-cross detecting circuit 80 detects each zero-cross point (i.e.,zero degree phase) in the exciting primary A.C. signal sinωt, so as togenerate a zero-cross detection pulse ZP. A phase difference detectingcircuit 81 outputs a gate pulse having a time width that corresponds toa generation timing difference +θ between a zero-cross detection pulse(latch pulse) LP1 of the output signal Y1(=sin(ωt+θ)) and the zero-crossdetection pulse ZP (more particularly, plus θ±d). This gate pulse isthen given to a voltage converting circuit 83, which in turn outputs anintegrated voltage +Vθ corresponding to the pulse time width (i.e., ananalog voltage corresponding to the phase amount+θ±d).

Another phase difference detecting circuit 82 outputs a gate pulsehaving a time width that corresponds to a generation timing difference−θ between the zero-cross detection pulse ZP and a zero-cross detectionpulse (latch pulse) LP2 of the output signal Y2(=sin(ωt−θ)) (moreparticularly, minus θ±d). This gate pulse is then given to a voltageconverting circuit 84, which in turn outputs an integrated voltage −Vθcorresponding to the pulse time width (i.e., an analog voltagecorresponding to the phase amount −θ±d). These voltages +Vθ and −Vθ areadded together by an adder 85, the resultant sum is divided by two via adivider 86, and then the quotient from the divider 86 is subtracted fromthe integrated voltage +Vθ. In this manner, these analog operatorsexecute arithmetic operations similar to those provided by the operators49 to 52 of FIG. 3, and consequently they can yield analog positiondetection data θ.

The circuitry illustrated in FIG. 8A may be simplified in a manner asshown in FIG. 8B, in which a phase difference detecting circuit 88outputs a gate pulse having a time width that corresponds to ageneration timing difference 2θ between the zero-cross detection pulse(latch pulse) LP1 of the output signal Y1(=sin(ωt+θ)) and the zero-crossdetection pulse (latch pulse) LP2 of the output signal Y2(=sin(ωt−θ)).This gate pulse is then given to a voltage converting circuit 89, whichin turn outputs an integrated voltage corresponding to the pulse timewidth (i.e., an analog voltage corresponding to the phase amount 2θ).The thus-determined analog voltage, which is a voltage having removedtherefrom an error ±d caused by temperature change etc., corresponds to(or is proportional to) θ and therefore can be utilized directly asposition detection data θ.

The above-described various embodiments are capable of detecting, in anabsolute value, a linear position x within a range of one pitch length pbetween the magnetic response members 22. Absolute values of linearpositions x beyond the pitch length p can be detected by an appropriatecounter incrementally or decrementally counting the number of occurringpitch lengths each time the object of detection moves beyond one of thepitch lengths p. This counting may be effected by incrementing ordecrementing the counted value of the counter by one depending on thedirection of movement of the magnetic response members 22, each time theoutput signal of the winding section 10 makes a round through theone-pitch-length range. For example, circuitry as shown in FIG. 9 may beprovided in such a manner that determining circuits 70 and 71 determinewhen the digital measurement value based on the output signal of thewinding section 10 changes from its maximum to minimum (M→0) or from itsminimum to maximum (0→M), so as to generate a count trigger signal of avalue “+1” or “−1” to be counted by a counter 72. In this case, thecount Np of the counter 72 can be used as higher-order data of aposition detection value.

Alternatively, two detecting sections differing from each other in onepitch length p may be provided on both sides of a single rod 210, asshown in FIGS. 10A and 10B, so that absolute values of linear positionsx beyond the pitch length p are detected on the basis of the vernierprinciple. FIG. 10A is an axial sectional view of the rod 210 takenalong the axis thereof, while FIG. 10B is a radial sectional view of therod 210 taken across the diameter thereof. The first detecting sectionfunctioning as a main measure has a plurality of recessed portions 21 c(or 21 d) formed therein in a repeated fashion along the length of amagnetic rod 210, so that a plurality of raised positions are formed asmagnetic response members 22 repeated at a predetermined pitch P1,thereby resulting in an alternating repetition of the recessed andraised portions 21 c (or 21 d) and 22. The winding section 10-1corresponding to the first detecting section includes four poles 11 to14 corresponding to the sine phase (s), cosine phase (c), minus sinephase (/s) and minus cosine phase (/c), respectively. Similarly, thesecond detecting section functioning as a secondary measure has aplurality of recessed portions 21 c′ formed therein in a repeatedfashion along the length of a magnetic rod 210, so that a plurality ofraised positions are formed as magnetic response members 22′ inrepetition at a predetermined pitch P2, thereby resulting in analternating repetition of the recessed and raised portions 21 c′ and22′. The winding section 10-2 corresponding to the second detectingsection includes four poles 11 to 14 corresponding to the sine phase(s), cosine phase (c), minus sine phase (/s) and minus cosine phase(/c), respectively. The pitches P1 and P2 in the first and seconddetecting sections differ from each other by an appropriate amount. Byarithmetically processing position detection data θ1 and θ2 from thefirst and second detecting sections, absolute position detection valuescan be obtained within a range of the least common multiple of the pitchlengths P1 and P2. In this case, the rod 210 is of course properlyguided to just linearly move without being accidentally rotated at all.

In each of the above-described embodiments, the known pulse-widthmodulation technique may be employed when position detection data θindicative of a position detected in an analog or digital manner is tobe transmitted via the wirings or lines to another device utilizing thedetection data θ (“utilizing device”). FIG. 11A shows a modifiedembodiment where analog position detection data θ output from an analogphase detection circuit, similar to that of FIG. 8A or 8B, is modulatedin pulse width via a pulse-width modulation circuit 100. For example,the pulse-width modulation circuit 100 includes an analog comparator 101and an analog triangular-wave generator circuit 102. FIG. 11B is adiagram showing how the pulse-width modulation circuit 100 performs thepulse width modulation. Namely, the analog comparator 101 makes acomparison between the analog position detection data θ and an analogtriangular-wave signal TRW generated by the analog triangular-wavegenerator circuit 102 and thereby outputs a pulse-width-modulatedposition detection signal PWMθ having a pulse width corresponding to avoltage level or value of the detection data θ. Thispulse-width-modulated position detection signal PWMθ is supplied viawirings 103 to a utilizing device 104, which utilizes the suppliedposition detection signal PWMθ in a desired manner. For instance, theutilizing device 104 may reproduce, from the pulse-width-modulatedposition detection signal PWMθ, the position detection data θ as ananalog or digital value. FIG. 11C is a timing chart explanatory of anexemplary manner in which the position detection data θ is reproduced bythe utilizing device 104. From the pulse-width-modulated positiondetection signal PWMθ, data Tθ indicative of the pulse width of thesignal PWMθ is created by digital count or analog integration. Thethus-created data Tθ may be utilized directly as the position detectiondata θ; however, to compensate for any possible error due to a variationin the period of the analog triangular-wave signal TRW resulting from atemperature change and the like, it is more preferable that a pulseperiod Tr of the pulse-width-modulated position detection signal PWMθ bedetermined through the digital count or analog integration to therebyprovide a ratio of the pulse period Tr to the data Tθ. For example,assuming that TR/2 corresponds to a 360° phase of the position detectiondata θ, the position detection data θ can be reproduced by performing anarithmetic operation of “2Tθ/Tr”.

It should be appreciated that the pulse-width modulation circuit 100 mayperform the pulse width modulation using any other suitable circuit thanthe triangular-wave generator circuit 102, such as an analogsawtooth-wave generator circuit. Further, of course, the period of thepulse-width-modulating analog triangular-wave signal TRW may be set tohave no relation whatsoever to an exciting A.C. signal sinωt, and it maybe set in any desired manner. Transmitting the position detection data θafter having been subjected to the pulse width modulation, as in theexample of FIG. 11A, is advantageous in the following respect. Namely,where the utilizing device 104 is considerably remote from the sensor,the transmission wirings 103 have to have an increased length, whichwould undesirably result in analog voltage level variations due toimpedance variations by the influence of wiring capacity, noise,temperature changes, etc. However, the use of the pulse-width-modulatedposition detection signal PWMθ in the modified example can reliablyavoid adverse effects of such analog voltage level variations, therebyconstantly guaranteeing a high detection accuracy.

Further, FIGS. 12A and 12B show several other modified examples whichare designed to modulate the pulse width of digital position detectiondata θ (or D) output from a digital phase detection circuit as shown inFIG. 3, 5 or 6. In the example of FIG. 12A, the digital positiondetection data θ (or D) is first converted into an analog signal, andthe thus-converted analog signal is fed to an analog-type pulse widthmodulation circuit 100, similar to the one of FIG. 11A, so as to providea pulse-width-modulated position detection signal PWMθ. Further, in theexample of FIG. 12B, the digital position detection data θ (or D) is fedto a digital-type pulse width modulation circuit 106 so as to provide apulse-width-modulated position detection signal PWMθ. The digital-typepulse width modulation circuit 106 comprises, for example, digitalversions of the comparator 101 and triangular-wave generator circuit 102shown in FIG. 11A, and the pulse width modulating operation in theexample of FIG. 12B may be substantially the same as in the example ofFIG. 11B. Transmitting the digital position detection data θ (or D)after having been subjected to pulse width modulation as in the examplesof FIGS. 12A and 12B is advantageous in that the necessary number of thewirings 103 for transmitting the pulse-width-modulated positiondetection signal PWMθ can be far smaller than the necessary number ofthe wirings for transmitting the digital data in a parallel fashion. Itis also advantageous in that it can eliminate the need forparallel-to-serial conversion of the digital data at a transmitting end,serial-to-parallel conversion of the digital data at a receiving end andsynchronizing between the parallel-to-serial conversion and theserial-to-parallel conversion and thereby reduce the costs as comparedto the case where the digital data is transmitted in a serial fashion.

FIG. 13 shows still another modified example which employs a phasedifference detection circuit 88 similar to that shown in FIG. 8B andwhere zero-cross detection pulses LP1 and LP2 output from the twozero-cross detection circuits 47 and 48 of FIG. 3 are fed to the phasedifference detection circuit 88 so that a gate pulse, having a timewidth corresponding to a difference between the times of generation ofthe zero-cross detection pulses, is generated and output directly as apulse-width-modulated position detection signal PWMθ. By transmittingthe pulse-width-modulated position detection signal PWMθ, this examplecan dispense with the related circuits and thereby significantly reducethe necessary costs while still affording the benefit of providingposition detection data whose errors due to temperature drift etc. havebeen appropriately compensated for as in the example of FIG. 8B.

The present invention should not be construed as limited only to theabove-described embodiments, and may be modified in a variety of ways.Further, although the arithmetic operations according to the presentinvention can of course be carried out via a hardware device based onhard-wired logic using an IC, LSI, gate arrays or a group of otherdiscrete circuits, the present invention is not limited to sucharrangements. For example, a software program may be built to performfunctions equal to the arithmetic and other operations as describedabove in relation to the embodiments of the present invention, and thisprogram may be executed by a computer, microprocessor or DSP (DigitalSignal Processor). Circuits recited in the claims of the presentapplication should be construed as embracing equivalent circuitfunctions that are implemented by a computer or electronic circuit groupwithin a processor executing the software program. In addition,implementing the claimed invention via a hybrid combination of ahardware device or circuits having fixed functions andsoftware-processed circuit functions is also within the scope of thepresent invention.

What is claimed is:
 1. A phase difference detection device for aposition detector, said position detector being excited by apredetermined reference signal to generate first and second A.C. outputsignals, said first A C. output signal having been amplitude-modulatedusing, as an amplitude coefficient, a first function value correspondingto a position-to-be-detected, and said second A.C. output signal havingbeen amplitude-modulated using, as an amplitude coefficient, a secondfunction value corresponding to the position-to-be-detected, said phasedifference detection device comprising: a phase shift circuitoperatively coupled to said position detector to shift an electric phaseof said received first A.C. output signal by a predetermined angle; afirst circuit operatively coupled to said phase shift circuit and saidposition detector to perform an operation between an output signal ofsaid phase shift circuit and said second A.C. output signal so as tosynthesize a first data signal having an electric phase angle shifted inone of positive and negative directions in correspondence to theposition-to-be-detected; a second circuit operatively coupled to saidphase shift circuit and said position detector to perform an operationbetween an output signal of said phase shift circuit and said secondA.C. output signal so as to synthesize a second data signal having anelectric phase angle shifted in other of positive and negativedirections in correspondence to the position-to-be-detected; a firstoperation circuit operatively coupled to said first circuit to measurean electric phase difference between said predetermined reference signaland said first data signal to obtain first phase data; a secondoperation circuit operatively coupled to said second circuit to measurean electric phase difference between said predetermined reference signaland said second data signal to obtain second phase data; a thirdoperation circuit operatively coupled to said first and second operationcircuit to calculate position detection data corresponding to theposition-to-be-detected on the basis of said first and second phasedata; and a pulse-width modulation circuit coupled to said thirdoperation circuit to generate a signal pulse-width-modulated inaccordance with the position detection data.
 2. A phase differencedetection device as claimed in claim 1 wherein said third operationcircuit calculates the position detection data as analog positiondetection data, and said pulse-width modulation circuit includes ananalog circuit for processing the analog position detection data.
 3. Aphase difference detection device as claimed in claim 1 wherein saidthird operation circuit calculates the position detection data asdigital position detection data, and said pulse-width modulation circuitincludes a digital circuit for processing the digital position detectiondata.
 4. A phase difference detection device as claimed in claim 1wherein said third operation circuit calculates the position detectiondata as digital position detection data, said phase difference detectiondevice further comprising a converter for converting the digitalposition detection data into analog position detection data, and whereinsaid pulse-width modulation circuit includes an analog circuit forprocessing the analog position detection data.
 5. A phase differencedetection device for a position detector, said position detector beingexcited by a predetermined reference signal to generate first and secondA.C. output signals, said first A C. output signal having beenamplitude-modulated using, as an amplitude coefficient, a first functionvalue corresponding to a position-to-be-detected, and said second A.C.output signal having been amplitude-modulated using, as an amplitudecoefficient, a second function value corresponding to theposition-to-be-detected, said phase difference detection devicecomprising: a phase shift circuit operatively coupled to said positiondetector to shift an electric phase of said received first A.C. outputsignal by a predetermined angle; a first circuit operatively coupled tosaid phase shift circuit and said position detector to perform anoperation between an output signal of said phase shift circuit and saidsecond A.C. output signal so as to synthesize a first data signal havingan electric phase angle shifted in one of positive and negativedirections in correspondence to the position-to-be-detected; a secondcircuit operatively coupled to said phase shift circuit and saidposition detector to perform an operation between an output signal ofsaid phase shift circuit and said second A.C. output signal so as tosynthesize a second data signal having an electric phase angle shiftedin other of positive and negative directions in correspondence to theposition-to-be-detected; and a third circuit operatively coupled to saidfirst and second circuit to generate, on the basis of a differencebetween said first data signal and said second data signal, a signalpulse-width-modulated in accordance with position data indicative of theposition-to-be-detected.
 6. A method of detecting a position by use of aposition detector, said position detector being excited by apredetermined reference signal to generate first and second outputsignals, said first output signal having been amplitude-modulated using,as an amplitude coefficient, a first function value corresponding to aposition-to-be-detected, and said second output signal having beenamplitude-modulated using, as an amplitude coefficient, a secondfunction value corresponding to the position-to-be-detected, said methodcomprising the steps of: receiving said first and second output signalsfrom said position detector; forming first and second data signals fromsaid received first and second output signals, said first data signalhaving an electric phase angle shifted in a positive direction incorrespondence to a position-to-be-detected, said second data signalhaving an electric phase angle shifted in a negative direction incorrespondence to said position-to-be-detected; measuring an electricphase difference between said predetermined reference signal and saidfirst data signal to obtain first phase data; measuring an electricphase difference between said predetermined reference signal and saidsecond data signal to obtain second phase data; operating positiondetection data corresponding to the position-to-be-detected on the basisof said first and second phase data; and generating a signalpulse-width-modulated in accordance with the position detection data. 7.A method of detecting a position by use of a position detector, saidposition detector being excited by a predetermined reference signal togenerate first and second output signals, said first output signalhaving been amplitude-modulated using, as an amplitude coefficient, afirst function value corresponding to a position-to-be-detected, andsaid second output signal having been amplitude-modulated using, as anamplitude coefficient, a second function value corresponding to theposition-to-be-detected, said method comprising the steps of: receivingsaid first and second output signals from said position detector;forming first and second data signals from said received first andsecond output signals, said first data signal having an electric phaseangle shifted in a positive direction in correspondence to aposition-to-be-detected, said second data signal having an electricphase angle shifted in a negative direction in correspondence to saidposition-to-be-detected; on the basis of a difference between said firstdata signal and said second data signal, generating a signalpulse-width-modulated in accordance with position data indicative of theposition-to-be-detected.